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1 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas monolithic 1.5a, 1.25mhz step-down switching regulators n 1.5a switch in a small msop package n constant 1.25mhz switching frequency n high power exposed pad (ms8e) package n wide operating voltage range: 3v to 25v n high efficiency 0.22 w switch n 1.2v feedback reference voltage n fixed output voltages of 1.8v, 2.5v, 3.3v, 5v n 2% overall output tolerance n uses low profile surface mount components n low shutdown current: 6 m a n synchronizable to 2mhz n current mode loop control n constant maximum switch current rating at all duty cycles* the lt ? 1767 is a 1.25mhz monolithic buck switching regulator. a high efficiency 1.5a, 0.22 w switch is included on the die together with all the control circuitry required to complete a high frequency, current mode switching regu- lator. current mode control provides fast transient re- sponse and excellent loop stability. new design techniques achieve high efficiency at high switching frequencies over a wide operating range. a low dropout internal regulator maintains consistent perfor- mance over a wide range of inputs from 24v systems to li- ion batteries. an operating supply current of 1ma im- proves efficiency, especially at lower output currents. shutdown reduces quiescent current to 6 m a. maximum switch current remains constant at all duty cycles. syn- chronization allows an external logic level signal to in- crease the internal oscillator from 1.4mhz to 2mhz. the lt1767 is available in an 8-pin msop fused leadframe package and a low thermal resistance exposed pad pack- age. full cycle-by-cycle short-circuit protection and ther- mal shutdown are provided. high frequency operation allows the reduction of input and output filtering compo- nents and permits the use of chip inductors. n dsl modems n portable computers n wall adapters n battery-powered systems n distributed power , ltc and lt are registered trademarks of linear technology corporation. efficiency vs load current 12v to 3.3v step-down converter boost lt1767-3.3 v in output 3.3v 1.2a* v in 12v 1767 ta01 c2 0.1 m f c c 1.5nf r c 4.7k d1 ups120 c1 10 m f ceramic c3 2.2 m f ceramic d2 cmdsh-3 l1 5 m h v sw fb shdn open or high = on gnd v c sync *maximum output current is subject to thermal derating. load current (a) 0 0.2 0.4 0.6 0.8 1 1.2 1.4 efficiency (%) 1767 ta01a 95 90 85 80 75 70 v in = 10v v out = 5v v in = 5v v out = 3.3v *patent pending applicatio s u features typical applicatio u descriptio u
2 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas absolute m axi m u m ratings w ww u input voltage .......................................................... 25v boost pin above sw ............................................ 20v max boost pin voltage .......................................... 35v shdn pin ............................................................... 25v fb pin voltage .......................................................... 6v fb pin current ....................................................... 1ma electrical characteristics (note 1) parameter condition min typ max units maximum switch current limit t a = 0 c to 125 c 1.5 2 3 a t a = < 0 c 1.3 3 a oscillator frequency 3.3v < v in < 25v 1.1 1.25 1.4 mhz l 1.1 1.5 mhz switch on voltage drop i sw = C1.5a, 0 c t a 125 c and C1.3a, t a < 0 c 330 400 mv l 500 mv v in undervoltage lockout (note 3) l 2.47 2.6 2.73 v v in supply current v fb = v nom + 17% l 1 1.3 ma shutdown supply current v shdn = 0v, v in = 25v, v sw = 0v 6 20 m a l 45 m a feedback voltage 3v < v in < 25v, 0.4v < v c < 0.9v lt1767 (adj) 1.182 1.2 1.218 v (note 3) l 1.176 1.224 v lt1767-1.8 l 1.764 1.8 1.836 v lt1767-2.5 l 2.45 2.5 2.55 v lt1767-3.3 l 3.234 3.3 3.366 v lt1767-5 l 4.9 5 5.1 v fb input current lt1767 (adj) l C 0.25 C 0.5 m a the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 15v, v c = 0.8v, boost = v in + 5v, shdn, sync and switch open unless otherwise noted. sync pin current .................................................. 1ma operating junction temperature range (note 2) lt1767e .......................................... C 40 c to 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c package/order i n for m atio n w u u order part number 1 2 3 4 boost v in sw gnd 8 7 6 5 sync v c fb shdn top view ms8 package 8-lead plastic msop t jmax = 125 c, q ja = 110 c/w ltls ltwg ltwd ltwe ltwf lt1767ems8 lt1767ems8-1.8 lt1767ems8-2.5 lt1767ems8-3.3 lt1767ems8-5 ground pin connected to large copper area ms8 part marking order part number t jmax = 125 c, q ja = 40 c/w lt1767ems8e lt1767ems8e-1.8 lt1767ems8e-2.5 lt1767ems8e-3.3 lt1767ems8e-5 exposed gnd pad connected to large copper area on pcb ms8e part marking ltzg ltzh ltzj ltzk ltzl 1 2 3 4 boost v in sw gnd 8 7 6 5 sync v c fb shdn top view ms8e package 8-lead plastic msop consult ltc marketing for parts specified with wider operating temperature ranges.
3 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas electrical characteristics fb input resistance lt1767-1.8 l 10.5 15 21 k w lt1767-2.5 l 14.7 21 30 k w lt1767-3.3 l 19 27.5 39 k w lt1767-5 l 29 42 60 k w error amp voltage gain 0.4v < v c < 0.9v 150 350 error amp transconductance d i vc = 10 m a l 500 850 1300 m mho v c pin source current v fb = v nom C 17% l 80 120 160 m a v c pin sink current v fb = v nom + 17% l 70 110 180 m a v c pin to switch current transconductance 2.5 a/v v c pin minimum switching threshold duty cycle = 0% 0.35 v v c pin 1.5a i sw threshold 0.9 v maximum switch duty cycle v c = 1.2v, i sw = 400ma 85 90 % l 80 % minimum boost voltage above switch i sw = C1.5a, 0 c t a 125 c and C1.3a, t a < 0 c l 1.8 2.7 v boost current i sw = C 0.5a (note 4) l 10 15 ma i sw = C1.5a, 0 c t a 125 c and C1.3a, t a < 0 c (note 4) l 30 45 ma shdn threshold voltage l 1.27 1.33 1.40 v shdn input current (shutting down) shdn = 60mv above threshold l C7 C10 C13 m a shdn threshold current hysteresis shdn = 100mv below threshold l 4710 m a sync threshold voltage 1.5 2.2 v sync input frequency 1.5 2 mhz sync pin resistance i sync = 1ma 20 k w the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 15v, v c = 0.8v, boost = v in + 5v, shdn, sync and switch open unless otherwise noted. parameter condition min typ max units note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: the lt1767e is guaranteed to meet performance specifications from 0 c to 125 c. specifications over the C 40 c to 125 c operating junction temperature range are assured by design, characterization and correlation with statistical process controls. note 3: minimum input voltage is defined as the voltage where the internal regulator enters lockout. actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. see applications information. note 4: current flows into the boost pin only during the on period of the switch cycle. typical perfor m a n ce characteristics uw temperature ( c) ?0 ?5 0 25 50 75 100 125 fb voltage (v) 1767 g01 1.22 1.21 1.20 1.19 1.18 switch current (a) 0 0.5 1 1.5 switch voltage (mv) 1767 g02 400 350 300 250 200 150 100 50 0 125 c 25 c ?0 c fb vs temperature (adj) switch on voltage drop oscillator frequency temperature ( c) ?0 ?5 0 25 50 75 100 125 frequency (mhz) 1767 g03 1.50 1.45 1.40 1.35 1.30 1.25 1.20 1.15 1.10
4 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas temperature ( c) ?0 ?5 0 25 50 75 100 125 shdn threshold (v) 1767 g04 1.40 1.38 1.36 1.34 1.32 1.30 v in (v) 0 5 10 15 20 25 30 v in current ( a) 1767 g05 7 6 5 4 3 2 1 0 shdn = 0v load current (a) 0.001 0.01 0.1 1 input voltage (v) 1767 g07 3.5 3.3 3.1 2.9 2.7 2.5 shutdown voltage (v) 0 0.2 0.4 0.6 0.8 1 1.2 1.4 v in current ( a) 1767 g08 300 250 200 150 100 50 0 v in = 15v input voltage (v) 0 5 10 15 20 25 30 v in current ( a) 1767 g09 1200 1000 800 600 400 200 0 minimum input voltage shdn i p current vs temperature shdn threshold vs temperature shdn supply current vs v in minimum input voltage for 2.5v out shdn supply current input supply current feedback voltage (v) 0 0.2 0.4 0.6 0.8 1 1.2 switch peak current (a) 1767 g10 2.0 1.5 1.0 0.5 0 fb input current ( a) 40 30 20 10 0 fb current switch current 1767 g11 input voltage (v) 0 5 10 15 20 25 output current (a) 1.5 1.3 1.1 0.9 0.7 0.5 l = 4.7 h l = 2.2 h l = 1.5 h input voltage (v) 05 10 15 20 25 output current (a) 1767 g12 1.5 1.3 1.1 0.9 0.7 l = 4.7 h l = 2.2 h l = 1.5 h current limit foldback maximum load current, v out = 5v maximum load current, v out = 2.5v typical perfor a ce characteristics uw temperature ( c) ?0 ?5 0 25 50 75 100 125 shdn input ( a) 1767 g06 ?2 ?0 ? ? ? ? 0 starting up shutting down
5 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas pi n fu n ctio n s uuu fb: the feedback pin is used to set output voltage using an external voltage divider that generates 1.2v at the pin with the desired output voltage. the fixed voltage 1.8v, 2.5v, 3.3v and 5v versions have the divider network included internally and the fb pin is connected directly to the output. if required, the current limit can be reduced during start up or short-circuit when the fb pin is below 0.5v (see the current limit foldback graph in the typical perfor- mance characteristics section). an impedance of less than 5k w (adjustable part only) at the fb pin is needed for this feature to operate. boost: the boost pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar npn power switch. without this added voltage, the typical switch voltage loss would be about 1.5v. the additional boost voltage allows the switch to saturate and voltage loss approximates that of a 0.22 w fet structure. v in : this is the collector of the on-chip power npn switch. this pin powers the internal circuitry and internal regula- tor. at npn switch on and off, high di/dt edges occur on this pin. keep the external bypass capacitor and catch diode close to this pin. all trace inductance in this path will create a voltage spike at switch off, adding to the v ce voltage across the internal npn. gnd: the gnd pin acts as the reference for the regulated output, so load regulation will suffer if the ground end of the load is not at the same voltage as the gnd pin of the ic. this condition will occur when load current or other currents flow through metal paths between the gnd pin and the load ground point. keep the ground path short between the gnd pin and the load and use a ground plane when possible. keep the path between the input bypass and the gnd pin short. the gnd pin of the ms8 package is directly attached to the internal tab. this pin should be attached to a large copper area to improve thermal resistance. the exposed pad of the ms8e package is also connected to gnd. this should be soldered to a large copper area to improve its thermal resistance. v sw : the switch pin is the emitter of the on-chip power npn switch. this pin is driven up to the input pin voltage during switch on time. inductor current drives the switch pin negative during switch off time. negative voltage must be clamped with an external catch diode with a v br <0.8v. sync: the sync pin is used to synchronize the internal oscillator to an external signal. it is directly logic compat- ible and can be driven with any signal between 20% and 80% duty cycle. the synchronizing range is equal to initial operating frequency, up to 2mhz. see synchronization section in applications information for details. when not in use, this pin should be grounded. shdn: the shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. the 1.33v threshold can function as an accurate under- voltage lockout (uvlo), preventing the regulator from operating until the input voltage has reached a predeter- mined level. float or pull high to put the regulator in the operating mode. v c : the v c pin is the output of the error amplifier and the input of the peak switch current comparator. it is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. this pin sits at about 0.35v for very light loads and 0.9v at maximum load. it can be driven to ground to shut off the output.
6 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas block diagra m w and output capacitor, then an abrupt 180 shift will occur. the current fed system will have 90 phase shift at a much lower frequency, but will not have the additional 90 shift until well beyond the lc resonant frequency. this makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. high switch efficiency is attained by using the boost pin to provide a voltage to the switch driver which is higher than the input voltage, allowing switch to be saturated. this boosted voltage is generated with an external capaci- tor and diode. a comparator connected to the shutdown pin disables the internal regulator, reducing supply current. the lt1767 is a constant frequency, current mode buck converter. this means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. in addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. a switch cycle starts with an oscilla- tor pulse which sets the r s flip-flop to turn the switch on. when switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. this technique means that the error amplifier commands current to be delivered to the output rather than voltage. a voltage fed system will have low phase shift up to the resonant frequency of the inductor figure 1. block diagram + + s v in 2.5v bias regulator 1.25mhz oscillator v sw fb v c gnd 1767 f01 slope comp 0.01 w internal v cc current sense amplifier voltage gain = 40 sync shdn shutdown comparator current comparator error amplifier g m = 850 mho boost r s flip-flop driver circuitry s r 0.35v q1 power switch parasitic diodes do not forward bias 1.2v + + 1.33v 3 a 7 a 2 8 5 7 1 4 6 3
7 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u fb resistor network if an output voltage of 1.8v, 2.5v, 3.3v or 5v is required, the respective fixed option part, -1.8, -2.5, -3.3 or -5, should be used. the fb pin is tied directly to the output; the necessary resistive divider is already included on the part. for other voltage outputs, the adjustable part should be used and an external resistor divider added. the suggested resistor (r2) from fb to ground is 10k. this reduces the contribution of fb input bias current to output voltage to less than 0.25%. the formula for the resistor (r1) from v out to fb is: r rv ra out 1 212 12 2025 = - () -m . .(.) of capacitance is less important and has no significant effect on loop stability. if operation is required close to the minimum input required by the output of the lt1767, a larger value may be required. this is to prevent excessive ripple causing dips below the minimum operating voltage, resulting in erratic operation. if tantalum capacitors are used, values in the 22 m f to 470 m f range are generally needed to minimize esr and meet ripple current and surge ratings. care should be taken to ensure the ripple and surge ratings are not exceeded. the avx tps and kemet t495 series are surge rated. avx recommends derating capacitor operating voltage by 2:1 for high surge applications. output capacitor unlike the input capacitor, rms ripple current in the output capacitor is normally low enough that ripple current rating is not an issue. the current waveform is triangular, with an rms value given by: i vvv lfv ripple rms out in out in () = () - () ()()( ) 029 . the lt1767 will operate with both ceramic and tantalum output capacitors. ceramic capacitors are generally chosen for their small size, very low esr (effective series resis- tance), and good high frequency operation, reducing out- put ripple voltage. their low esr removes a useful zero in the loop frequency response, common to tantalum capaci- tors. to compensate for this, the v c loop compensation pole frequency must typically be reduced by a factor of 10. typical ceramic output capacitors are in the 1 m f to 10 m f range. since the absolute value of capacitance defines the pole frequency of the output stage, an x7r or x5r type ceramic, which have good temperature stability, is recom- mended. tantalum capacitors are usually chosen for their bulk capacitance properties, useful in high transient load appli- cations. esr rather than capacitive value defines output ripple at 1.25mhz. typical lt1767 applications require a tantalum capacitor with less than 0.3 w esr at 22 m f to 500 m f, see table 2. figure 2. feedback network + 1.2v v sw v c gnd 1767 f02 r1 r2 10k output error amplifier fb lt1767 + input capacitor step-down regulators draw current from the input supply in pulses. the rise and fall times of these pulses are very fast. the input capacitor is required to reduce the voltage ripple this causes at the input of lt1767 and force the switching current into a tight local loop, thereby minimizing emi. the rms ripple current can be calculated from: iivvvv ripple rms out out in out in () =- () / 2 higher value, lower cost ceramic capacitors are now avail- able in smaller case sizes. these are ideal for input bypass- ing since their high frequency capacitive nature removes most ripple current rating and turn-on surge problems. at higher switching frequency, the energy storage require- ment of the input capacitor is reduced so values in the range of 1 m f to 4.7 m f are suitable for most applications. y5v or similar type ceramics can be used since the absolute value
8 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u figure 3. output ripple voltage waveform and reduces the current at which discontinuous operation occurs. the following formula gives maximum output current for continuous mode operation, implying that the peak to peak ripple (2x the term on the right) is less than the maximum switch current. continuous mode i out max () = i p - () - () ()()( ) vvv lfv out in out in 2 discontinuous operation occurs when i v lf out dis out () () ()() = 2 for v in = 8v, v out = 5v and l = 3.3 m h, i a out max () - =- () - () ()() () =- = 15 58 5 2 3 3 10 1 25 10 8 15 023 127 66 . . . .. . note that the worst case (minimum output current avail- able) condition is at the maximum input voltage. for the same circuit at 15v, maximum output current would be only 1.1a. when choosing an inductor, consider maximum load current, core and copper losses, allowable component height, output voltage ripple, emi, fault current in the inductor, saturation, and of course, cost. the following procedure is suggested as a way of handling these some- what complicated and conflicting requirements. 1. choose a value in microhenries from the graphs of maximum load current. choosing a small inductor with lighter loads may result in discontinuous mode of operation, but the lt1767 is designed to work well in either mode. assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. if maxi- mum load current is 0.5a, for instance, a 0.5a inductor may not survive a continuous 2a overload condition. also, the instantaneous application of input or release from shutdown, at high input voltages, may cause inductor choice and maximum output current maximum output current for a buck converter is equal to the maximum switch rating (i p ) minus one half peak to peak inductor current. in past designs, the maximum switch current has been reduced by the introduction of slope compensation. slope compensation is required at duty cycles above 50% to prevent an affect called subharmonic oscillation (see application note 19 for details). the lt1767 has a new circuit technique that maintains a constant switch current rating at all duty cycles. (patent pending) for most applications, the output inductor will be in the 1 m h to 10 m h range. lower values are chosen to reduce the physical size of the inductor, higher values allow higher output currents due to reduced peak to peak ripple current, table 2. surface mount solid tantalum capacitor esr and ripple current e case size esr (max, w ) ripple current (a) avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.7 to 0.9 0.4 d case size avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 c case size avx tps 0.2 (typ) 0.5 (typ) figure 3 shows a comparison of output ripple for a ceramic and tantalum capacitor at 200ma ripple current. v out using 47 m f, 0.1 w tantalum capacitor (10mv/div) 0.2 m s/div 1767 f03 v out using 2.2 m f ceramic capacitor (10mv/div) v sw (5v/div)
9 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u 4. after making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. use the experts in the linear technologys applica- tions department if you feel uncertain about the final choice. they have experience with a wide range of inductor types and can tell you about the latest devel- opments in low profile, surface mounting, etc. catch diode the suggested catch diode (d1) is a ups120 schottky, or its motorola equivalent, mbrm120lti/mbrm130lti. it is rated at 2a average forward current and 20v/30v reverse voltage. typical forward voltage is 0.5v at 1a. the diode conducts current only during switch off time. peak reverse voltage is equal to regulator input voltage. average forward current in normal operation can be calculated from: i ivv v d avg out in out in () = - () boost pin for most applications, the boost components are a 0.1 m f capacitor and a cmdsh-3 diode. the anode is typically connected to the regulated output voltage to generate a voltage approximately v out above v in to drive the output stage. the output driver requires at least 2.7v of head- room throughout the on period to keep the switch fully saturated. however, the output stage discharges the boost capacitor during the on time. if the output voltage is less than 3.3v, it is recommended that an alternate boost supply is used. the boost diode can be connected to the input, although, care must be taken to prevent the 2x v in boost voltage from exceeding the boost pin absolute maximum rating. the additional voltage across the switch driver also increases power loss, reducing efficiency. if available, an independent supply can be used with a local bypass capacitor. a 0.1 m f boost capacitor is recommended for most appli- cations. almost any type of film or ceramic capacitor is saturation of the inductor. in these applications, the soft-start circuit shown in figure 10 should be used. 2. calculate peak inductor current at full load current to ensure that the inductor will not saturate. peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so dont omit this step. powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. other core materials fall somewhere in between. ii vvv lfv peak out out in out in =+ - () ()()( ) 2 v in = maximum input voltage f = switching frequency, 1.25mhz 3. decide if the design can tolerate an open core geom- etry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent emi problems. this is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. table 3 part number value (uh) i sat (amps) dcr ( w ) height (mm) coiltronics tp1-2r2 2.2 1.3 0.188 1.8 tp2-2r2 2.2 1.5 0.111 2.2 tp3-4r7 4.7 1.5 0.181 2.2 tp4- 100 10 1.5 0.146 3.0 murata lqh1c1r0m04 1.0 0.51 0.28 1.8 lqh3c1r0m24 1.0 1.0 0.06 2.0 lqh3c2r2m24 2.2 0.79 0.1 2.0 lqh4c1r5m04 1.5 1.0 0.09 2.6 sumida cd73- 100 10 1.44 0.080 3.5 cdrh4d18-2r2 2.2 1.32 0.058 1.8 cdrh5d18-6r2 6.2 1.4 0.071 1.8 cdrh5d28-100 10 1.3 0.048 2.8
10 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas suitable, but the esr should be <1 w to ensure it can be fully recharged during the off time of the switch. the capacitor value is derived from worst-case conditions of 700ns on-time, 50ma boost current, and 0.7v discharge ripple. this value is then guard banded by 2x for secondary factors such as capacitor tolerance, esr and temperature effects. the boost capacitor value could be reduced under less demanding conditions, but this will not improve circuit operation or efficiency. under low input voltage and low load conditions, a higher value capacitor will reduce discharge ripple and improve start up operation. shutdown and undervoltage lockout figure 4 shows how to add undervoltage lockout (uvlo) to the lt1767. typically, uvlo is used in situations where the input supply is current limited , or has a relatively high source resistance. a switching regulator draws constant power from the source, so source current increases as source voltage drops. this looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. uvlo prevents the regulator from operating at source voltages where these problems might occur. r vv a r v vv r a hl h 1 7 2 133 133 1 3 = - m = - () +m . . v h e turn-on threshold v l e turn-off threshold example: switching should not start until the input is above 4.75v and is to stop if the input falls below 3.75v. v h = 4.75v v l = 3.75v r vv a k r v vv k a k 1 475 375 7 143 2 133 475 133 143 3 49 4 = - m = = - () +m = .. . .. . keep the connections from the resistors to the shdn pin short and make sure that the interplane or surface capaci- tance to the switching nodes are minimized. if high resis- tor values are used, the shdn pin should be bypassed with a 1nf capacitor to prevent coupling problems from the switch node. synchronization the sync pin, is used to synchronize the internal oscilla- tor to an external signal. the sync input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 20% and 80%. the input can be driven directly from a logic level output. the synchronizing range is equal to initial operating frequency up to 2mhz. this means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (1.5mhz), not the typical operating frequency of 1.25mhz. caution should be used when synchronizing above 1.6mhz because at higher sync frequencies the amplitude of the internal slope compensation used to applicatio n s i n for m atio n wu u u 1.33v gnd v sw v in r1 1767 f04 output shdn v cc in lt1767 3 a r2 c1 + 7 a figure 4. undervoltage lockout an internal comparator will force the part into shutdown below the minimum v in of 2.6v. this feature can be used to prevent excessive discharge of battery-operated sys- tems. if an adjustable uvlo threshold is required, the shutdown pin can be used. the threshold voltage of the shutdown pin comparator is 1.33v. a 3 m a internal current source defaults the open pin condition to be operating (see typical performance graphs). current hysteresis is added above the shdn threshold. this can be used to set voltage hysteresis of the uvlo using the following:
11 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u prevent subharmonic switching is reduced. this type of subharmonic switching only occurs at input voltages less than twice output voltage. higher inductor values will tend to eliminate this problem. see frequency compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. application note 19 has more details on the theory of slope compensation. layout considerations as with all high frequency switchers, when considering layout, care must be taken in order to achieve optimal electrical, thermal and noise performance. for maximum efficiency, switch rise and fall times are typically in the nanosecond range. to prevent noise both radiated and conducted, the high speed switching current path, shown in figure 5, must be kept as short as possible. this is implemented in the suggested layout of figure 6. shorten- ing this path will also reduce the parasitic trace inductance of approximately 25nh/inch. at switch off, this parasitic inductance produces a flyback spike across the lt1767 switch. when operating at higher currents and input voltages, with poor layout, this spike can generate volt- ages across the lt1767 that may exceed its absolute maximum rating. a ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise. board layout also has a significant effect on thermal resistance. soldering the exposed pad to as large a copper area as possible and placing feedthroughs under the pad to a ground plane, will reduce die temperature and in- crease the power capacity of the lt1767. for the nonexposed package, pin 4 is connected directly to the pad inside the package. similar treatment of this pin will result in lower die temperatures. thermal calculations power dissipation in the lt1767 chip comes from four sources: switch dc loss, switch ac loss, boost circuit current, and input quiescent current. the following formulas show how to calculate each of these losses. these formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. switch loss: p ri v v ns i v f sw sw out out in out in = ()( ) + ()()() 2 17 boost current loss for v boost = v out : p vi v boost out out in = () 2 50 / quiescent current loss: pv qin = () 0 001 . r sw = switch resistance ( ? 0.27 w when hot) 17ns = equivalent switch current/voltage overlap time f = switch frequency example: with v in = 10v, v out = 5v and i out = 1a: p w pw pw sw boost q = ( )()() + () ()( ) () =+= = ()( ) = = () = - 027 1 5 10 17 10 1 10 1 25 10 0 135 0 21 0 34 5150 10 005 10 0 001 0 01 2 96 2 . . ... / . .. figure 5. high speed switching path 1767 f05 5v l1 sw v in lt1767 d1 c1 c3 v in high frequency circulating path load the v c and fb components should be kept as far away as possible from the switch and boost nodes. the lt1767 pinout has been designed to aid in this. the ground for these components should be separated from the switch current path. failure to do so will result in poor stability or subharmonic like oscillation.
12 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u total power dissipation is 0.34 + 0.05 + 0.01 = 0.4w. thermal resistance for lt1767 package is influenced by the presence of internal or backside planes. with a full plane under the package, thermal resistance for the exposed pad package will be about 40 c/w. no plane will increase resistance to about 150 c/w. to calculate die temperature, use the appropriate thermal resistance number and add in worst-case ambient temperature: t j = t a + q ja (p tot ) when estimating ambient, remember the nearby catch diode and inductor will also be dissipating power. p vv v i v diode f in out load in = () - ()() v f = forward voltage of diode (assume 0.5v at 1a) pw diode = () - ()() = 05 12 5 1 12 029 . . figure 6. typical application and suggested layout (topside only shown) boost lt1767-2.5 v in output 2.5v 1.2a v in 12v 1767 f06a c2 0.1 m f c c 1.5nf r c 4.7k d1 ups120 c1 10 m f ceramic c3 2.2 m f ceramic d2 cmdsh-3 l1 5 m h v sw fb shdn open or high = on gnd v c sync v in gnd r c c c v out c1 c3 c2 l1 1767 f06 sync shdn kelvin sense v out connect to ground plane minimize lt1767, c3, d1 loop keep fb and v c components away from high input components place feedthroughs around ground pin and under ground pad for good thermal conductivity d2 d1 gnd
13 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u notice that the catch diodes forward voltage contributes a significant loss in the overall system efficiency. a larger, lower v f diode can improve efficiency by several percent. p inductor = (i load ) (l dcr ) l dcr = inductor dc resistance (assume 0.1 w ) p inductor = (1) (0.1) = 0.1w typical thermal resistance of the board is 35 c/w. at an ambient temperature of 65 c, t j = 65 + 40 (0.4) + 35 (0.39) = 95 c if a true die temperature is required, a measurement of the sync to gnd pin resistance can be used. the sync pin resistance across temperature must first be calibrated, with no device power, in an oven. the same measurement can then be used in operation to indicate the die tempera- ture. frequency compensation before starting on the theoretical analysis of frequency response, the following should be remembered C the worse the board layout, the more difficult the circuit will be to stabilize. this is true of almost all high frequency analog circuits, read the layout considerations section first. common layout errors that appear as stability prob- lems are distant placement of input decoupling capacitor and/or catch diode, and connecting the v c compensation to a ground track carrying significant switch current. in addition, the theoretical analysis considers only first order non-ideal component behavior. for these reasons, it is important that a final stability check is made with produc- tion layout and components. the lt1767 uses current mode control. this alleviates many of the phase shift problems associated with the inductor. the basic regulator loop is shown in figure 7, with both tantalum and ceramic capacitor equivalent cir- cuits. the lt1767 can be considered as two g m blocks, the error amplifier and the power stage. figure 8 shows the overall loop response with a 330pf v c capacitor and a typical 100 m f tantalum output capacitor. the response is set by the following terms: error amplifier: dc gain set by g m and r l = 850 m ? 500k = 425. pole set by c f and r l = (2 p ? 500k ? 330p) C1 = 965hz. unity-gain set by c f and g m = (2 p ? 330p ? 850 m C1 ) C1 = 410khz. power stage: dc gain set by g m and r l (assume 10 w ) = 2.5 ? 10 = 25. pole set by c out and r l = (2 p ? 100 m ? 10) C1 = 159hz. unity-gain set by c out and g m = (2 p ? 100 m ? 2.5 C1 ) C1 = 3.98khz. tantalum output capacitor: zero set by c out and c esr = (2 p ? 100 m ? 0.1) C1 = 15.9khz. figure 8. overall loop response figure 7. model for loop response + 1.2v v sw v c lt1767 gnd 1767 f07 r1 output esr c f c c r c 500k error amplifier fb r2 c1 current mode power stage g m = 2.5mho g m = 850 mho + esl ceramic tantalum c1 frequency (hz) gain (db) 80 60 40 20 0 ?0 ?0 phase (deg) 180 150 120 90 60 30 0 1767 f10 gain phase v out = 5v c out = 100 f, 0.1 c c = 330pf r c /c f = n/c i load = 500ma 10 1k 10k 1m 100 100k
14 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u the zero produced by the esr of the tantalum output capacitor is very useful in maintaining stability. ceramic output capacitors do not have a zero due to very low esr, but are dominated by their esl. they form a notch in the 1mhz to 10mhz range. without this zero, the v c pole must be made dominant. a typical value of 2.2nf will achieve this. if better transient response is required, a zero can be added to the loop using a resistor (r c ) in series with the compensation capacitor. as the value of r c is increased, transient response will generally improve, but two effects limit its value. first, the combination of output capacitor esr and a large r c may stop loop gain rolling off alto- gether. second, if the loop gain is not rolled sufficiently at the switching frequency, output ripple will perturb the v c pin enough to cause unstable duty cycle switching similar to subharmonic oscillation. this may not be apparent at the output. small signal analysis will not show this since a continuous time system is assumed. if needed, an additional capacitor (c f ) can be added to form a pole at typically one fifth the switching frequency (if r c = ~ 5k, c f = ~ 100pf). when checking loop stability, the circuit should be oper- ated over the applications full voltage, current and tem- perature range. any transient loads should be applied and the output voltage monitored for a well-damped behavior. see application note 76 for more details. converter with backup output regulator in systems with a primary and backup supply, for ex- ample, a battery powered device with a wall adapter input, the output of the lt1767 can be held up by the backup supply with its input disconnected. in this condition, the sw pin will source current into the v in pin. if the shdn pin is held at ground, only the shut down current of 6 m a will be pulled via the sw pin from the second supply. with the shdn pin floating, the lt1767 will consume its quiescent operating current of 1ma. the v in pin will also source current to any other components connected to the input line. if this load is greater than 10ma or the input could be shorted to ground, a series schottky diode must be added, as shown in figure 9. with these safeguards, the output can be held at voltages up to the v in absolute maximum rating. buck converter with adjustable soft-start large capacitive loads or high input voltages can cause high input currents at start-up. figure 10 shows a circuit that limits the dv/dt of the output at start-up, controlling the capacitor charge rate. the buck converter is a typical configuration with the addition of r3, r4, c ss and q1. as the output starts to rise, q1 turns on, regulating switch current via the v c pin to maintain a constant dv/dt at the output. output rise time is controlled by the current through c ss defined by r4 and q1s v be . once the output is in regulation, q1 turns off and the circuit operates normally. r3 is transient protection for the base of q1. risetime rc v v ss out be = ()( )( ) () 4 using the values shown in figure 10, risetime ms == ( )( )() . 47 10 15 10 5 07 5 39 the ramp is linear and rise times in the order of 100ms are possible. since the circuit is voltage controlled, the ramp rate is unaffected by load characteristics and maximum output current is unchanged. variants of this circuit can be used for sequencing multiple regulator outputs. dual output sepic converter the circuit in figure 11 generates both positive and negative 5v outputs with a single piece of magnetics. the two inductors shown are actually just two windings on a standard b h electronics inductor. the topology for the 5v output is a standard buck converter. the C 5v topology would be a simple flyback winding coupled to the buck converter if c4 were not present. c4 creates a sepic (single-ended primary inductance converter) topology which improves regulation and reduces ripple current in l1. without c4, the voltage swing on l1b compared to l1a would vary due to relative loading and coupling losses. c4 provides a low impedance path to maintain an equal voltage swing in l1b, improving regulation. in a flyback converter, during switch on time, all the converters energy is stored in l1a only, since no current flows in l1b. at switch off, energy is transferred by magnetic coupling into l1b, powering the C 5v rail. c4 pulls l1b positive
15 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas applicatio n s i n for m atio n wu u u 3.3v, 1a * only required if input can sink >10ma removable input 0.1 m f 1.5nf 83k ups120 1767 f09 2.2 m f ups120* cmdsh-3 5 m h 2.2 f boost lt1767-3.3 v in v sw fb shdn gnd v c sync alternate supply 28.5k 4.7k figure 9. dual source supply with 6 m a reverse leakage figure 10. buck converter with adjustable soft-start boost lt1767-5 v in output 5v 1a v in 12v 1767 f10 c2 0.1 m f c1 100 m f c ss 15nf c c 330pf d1 c3 2.2 m f d2 cmdsh-3 l1 5 m h r3 2k v sw fb shdn gnd v c sync + + r4 47k q1 d1: ups120 q1: 2n3904 output 5v output ?v ? * l1 is a single core with two windings bh electronics #511-1013 ? if load can go to zero, an optional preload of 1k to 5k may be used to improve load regulation d1, d3: ups120 v in 6v to 15v gnd 1767 f11 c2 0.1 m f c c 330pf d1 c1 100 m f 10v tant c5 100 m f 10v tant c3 2.2 m f 16v ceramic c4 2.2 f 16v ceramic d2 cmdsh-3 d3 l1a* 9 m h l1b* + + boost lt1767-5 v in v sw fb shdn gnd v c sync figure 11. dual output sepic converter during switch on time, causing current to flow, and energy to build in l1b and c4. at switch off, the energy stored in both l1b and c4 supply the C5v rail. this reduces the information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. current in l1a and changes l1b current waveform from square to triangular. for details on this circuit, including maximum output currents, see design note 100. package descriptio n u msop (ms8) 1100 * dimension does not include mold flash, protrusions or gate burrs. mold flash, protrusions or gate burrs shall not exceed 0.006" (0.152mm) per side ** dimension does not include interlead flash or protrusions. interlead flash or protrusions shall not exceed 0.006" (0.152mm) per side 0.021 0.006 (0.53 0.015) 0 ?6 typ seating plane 0.007 (0.18) 0.043 (1.10) max 0.009 ?0.015 (0.22 ?0.38) 0.005 0.002 (0.13 0.05) 0.034 (0.86) ref 0.0256 (0.65) bsc 12 3 4 0.193 0.006 (4.90 0.15) 8 7 6 5 0.118 0.004* (3.00 0.102) 0.118 0.004** (3.00 0.102) ms8 package 8-lead plastic msop (reference ltc dwg # 05-08-1660)
16 lt1767/lt1767-1.8/ lt1767-2.5/lt1767-3.3/lt1767-5 sn1767 1767fas part number description comments lt1370 high efficiency dc/dc converter 42v, 6a, 500khz switch lt1371 high efficiency dc/dc converter 35v, 3a, 500khz switch lt1372/lt1377 500khz and 1mhz high efficiency 1.5a switching regulators boost topology lt1374 high efficiency step-down switching regulator 25v, 4.5a, 500khz switch lt1375/lt1376 1.5a step-down switching regulators 500khz, synchronizable in so-8 package lt1507 1.5a step-down switching regulator 500khz, 4v to 16v input, so-8 package lt1576 1.5a step-down switching regulator 200khz, reduced emi generation lt1578 1.5a step-down switching regulator 200khz, reduced emi generation lt1616 600ma step-down switching regulator 1.4mhz, 4v to 25v input, sot-23 package lt1676/lt1776 wide input range step-down switching regulators 60v input, 700ma internal switches ltc1765 1.25mhz, 3a wide input range step-down dc/dc v th = 3v to 25v, so-8 and tssop-16e packages ltc1877 high efficiency monolithic step-down regulator 550khz, ms8, v in up to 10v, i q =10 m a, i out to 600ma at v in = 5v ltc1878 high efficiency monolithic step-down regulator 550khz, ms8, v in up to 6v, i q = 10 m a, i out to 600ma at v in = 3.3v ltc3401 single cell, high current (1a), micropower, synchronous 3mhz v in = 0.5v to 5v, up to 97% efficiency synchronizable step-up dc/dc converter oscillator from 100khz to 3mhz ltc3402 single cell, high current (2a), micropower, synchronous 3mhz v in = 0.7v to 5v, up to 95% efficiency synchronizable step-up dc/dc converter oscillator from 100khz to 3mhz ltc3404 1.4mhz high efficiency, monolithic synchronous step-down up to 95% efficiency, 100% duty cycle, iq = 10 m a, regulator v in = 2.65v to 6v burst mode is a trademark of linear technology corporation. related parts package descriptio n u msop (ms8e) 0102 0.53 0.015 (.021 .006) seating plane note: 1. dimensions in millimeter/(inch) 2. drawing not to scale 3. dimension does not include mold flash, protrusions or gate burrs. mold flash, protrusions or gate burrs shall not exceed 0.152mm (.006") per side 4. dimension does not include interlead flash or protrusions. interlead flash or protrusions shall not exceed 0.152mm (.006") per side 5. lead coplanarity (bottom of leads after forming) shall be 0.102mm (.004") max 0.18 (.077) 0.254 (.010) 1.10 (.043) max 0.22 ?0.38 (.009 ?.015) 0.13 0.05 (.005 .002) 0.86 (.034) ref 0.65 (.0256) bcs 0 ?6 typ detail ? detail ? gauge plane 12 3 4 4.88 0.1 (.192 .004) 8 8 1 bottom view of exposed pad option 7 6 5 3.00 0.102 (.118 .004) (note 3) 3.00 0.102 (.118 .004) note 4 0.52 (.206) ref 1.83 0.102 (.072 .004) 2.06 0.102 (.080 .004) 5.23 (.206) min 3.2 ?3.45 (.126 ?.136) 2.083 0.102 (.082 .004) 2.794 0.102 (.110 .004) 0.889 0.127 (.035 .005) recommended solder pad layout 0.42 0.04 (.0165 .0015) typ 0.65 (.0256) bsc ms8e package 8-lead plastic msop (reference ltc dwg # 05-08-1662) lt/tp 0302 rev a 2k ? printed in usa linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear.com ? linear technology corporation 1999
step-down (buck) regulators home > products > power management > switching regulator > step-down (buck) regulators > internal power switch buck > lt1767 micropower buck internal power switch buck high input voltage buck poly-phase buck controllers external power switch buck controllers multiple output buck site help site map site index send us feedback ? 2007 linear technology | terms of use | privacy policy search lt1767 - monolithic 1.5a, 1.25mhz step-down switching regulators features click here for lt1767 evaluation kit 1.5a switch in a small msop-8 package constant 1.25mhz switching frequency high power exposed pad (ms8e) package wide operating voltage range: 3v to 25v high efficiency 0.22 ohm switch 1.2v feedback reference voltage fixed output voltages of 1.8v, 2.5v, 3.3v, 5v 2% overall output tolerance uses low profile surface mount components low shutdown current: 6a synchronizable to 2mhz current mode loop control constant maximum switch current rating at all duty cycles* typical application order now request samples documentation datasheet lt1767 - monolithi c 1.25mhz step - dow regulators lt magazine nov 2000 1.25mhz , monolithic step - do w keeps switching n o the signal spectru m lt chronicle oct 2002 power m a solutions for progr a logic ics sep 2002 industrial control reliability data r388 reliability da t reference design dc502 - lt1767 e v (pdf) software and simulatio n lt1767 demo circ u pa g e 1 of 5 linear technolo gy - lt1767 - monolithi c 1.5a, 1.25mhz ste p -down switchin g re g ulat... 18-se p -2008 htt p ://www.linear.com/ p c/ p roductdetail. j s p ?navid=h0,c1,c1003,c1042,c1032,c1064,...
description the lt1767 is a 1.25mhz monolithic buck switching regulator. a high efficiency 1.5a, 0.22 ohm switch is included on the die together with all the control circuitry required to complete a high frequency, current mode switching regulator. current mode contro l provides fast transient response and excellent loop stability. new design techniques achieve high efficiency at high switching frequencies over a wide operating range. a low dropout internal regulator maintains consistent performance ov er a wide range of inputs from 24v systems to li-ion batteries. an operat ing supply current of 1ma improves efficiency, especially at lower output currents. shutdown reduces quiescent current to 6a. maximum switch current remains constant at all duty cycles. synchronization allows an external logic level signal to increase the internal oscillator from 1.4mhz to 2mhz. the lt1767 is available in an 8-pin msop fused leadframe package and a low thermal resistance exposed pad package. full cycle-by-cycle short- circuit protection and thermal s hutdown are provided. high frequency operation allows the reduction of i nput and output filtering components and permits the use of chip inductors. packaging ms-8, ms-8e back to top back to top pa g e 2 of 5 linear technolo gy - lt1767 - monolithic 1.5a, 1.25mhz ste p -down switchin g re g ulat... 18-se p -2008 htt p ://www.linear.com/ p c/ p roductdetail. j s p ?navid=h0,c1,c1003,c1042,c1032,c1064,...
order info part numbers ending in pbf are lead free . please contact ltc marketing for information on lead based finish parts. part numbers containing tr or trm are shipped in tape and reel or 500 unit mini tape and reel , respectively please refer to our general ordering information or the product datasheet for more details package variations and pricing back to top part number package pins temp price (1-99) price (1k) * rohs data lt1767ems8 msop 8 e $3.45 $2.95 view lt1767ems8#pbf msop 8 e $3.45 $2.95 view lt1767ems8#tr msop 8 e $3.01 view lt1767ems8#trpbf msop 8 e $3.01 view pa g e 3 of 5 linear technolo gy - lt1767 - monolithic 1.5a, 1.25mhz ste p -down switchin g re g ulat... 18-se p -2008 htt p ://www.linear.com/ p c/ p roductdetail. j s p ?navid=h0,c1,c1003,c1042,c1032,c1064,...
* the usa list pricing shown is for budgetary use only, shown in united states dollars (fob usa per unit for the stated volume), and is subject to change. international prices may differ due to local duties, taxes, fees and exchange rates. for volume-specific price or delivery quotes, please contact your local linear technology sales office or authorized lt1767ems8-1.8 msop 8 e $3.45 $2.95 view lt1767ems8-1.8#pbf msop 8 e $3.45 $2.95 view lt1767ems8-1.8#tr msop 8 e $3.01 view lt1767ems8- 1.8#trpbf msop 8 e $3.01 view lt1767ems8-2.5 msop 8 e $3.45 $2.95 view lt1767ems8-2.5#pbf msop 8 e $3.45 $2.95 view lt1767ems8-2.5#tr msop 8 e $3.01 view lt1767ems8- 2.5#trpbf msop 8 e $3.01 view lt1767ems8-3.3 msop 8 e $3.45 $2.95 view lt1767ems8-3.3#pbf msop 8 e $3.45 $2.95 view lt1767ems8-3.3#tr msop 8 e $3.01 view lt1767ems8- 3.3#trpbf msop 8 e $3.01 view lt1767ems8-5 msop 8 e $3.45 $2.95 view lt1767ems8-5#pbf msop 8 e $3.45 $2.95 view lt1767ems8-5#tr msop 8 e $3.01 view lt1767ems8-5#trpbf msop 8 e $3.01 view lt1767ems8e msop 8 e $4.36 $3.05 view lt1767ems8e#pbf msop 8 e $4.36 $3.05 view lt1767ems8e#tr msop 8 e $3.11 view lt1767ems8e#trpbf msop 8 e $3.11 view lt1767ems8e-1.8 msop 8 e $4.36 $3.05 view lt1767ems8e-1.8#pbf msop 8 e $4.36 $3.05 view lt1767ems8e-1.8#tr msop 8 e $3.11 view lt1767ems8e- 1.8#trpbf msop 8 e $3.11 view lt1767ems8e-2.5 msop 8 e $4.36 $3.05 view lt1767ems8e-2.5#pbf msop 8 e $4.36 $3.05 view lt1767ems8e-2.5#tr msop 8 e $3.11 view lt1767ems8e- 2.5#trpbf msop 8 e $3.11 view lt1767ems8e-3.3 msop 8 e $4.36 $3.05 view lt1767ems8e-3.3#pbf msop 8 e $4.36 $3.05 view lt1767ems8e-3.3#tr msop 8 e $3.11 view lt1767ems8e- 3.3#trpbf msop 8 e $3.11 view lt1767ems8e-5 msop 8 e $4.36 $3.05 view lt1767ems8e-5#pbf msop 8 e $4.36 $3.05 view lt1767ems8e-5#tr msop 8 e $3.11 view lt1767ems8e- 5#trpbf msop 8 e $3.11 view buy now request samples pa g e 4 of 5 linear technolo gy - lt1767 - monolithic 1.5a, 1.25mhz ste p -down switchin g re g ulat... 18-se p -2008 htt p ://www.linear.com/ p c/ p roductdetail. j s p ?navid=h0,c1,c1003,c1042,c1032,c1064,...
distributor . evaluation kits applications dsl modems portable computers wall adapters battery-powered systems distributed power simulate ltspice / switchercad iii is a powerful free circuit simulator and schematic capture program, providing macro models for 80% of linear technology's switching regulators, over 200 op amp models, as well as resistors, transistors and mosfet models. step 1. if you do not already have a copy of ltspice, click here to download and install ltspice / switchercad iii step 2. once ltspice is installed, click here for a ready to run demonstration circuit of the lt1767 . step 3. if ltspice / switchercad i ii does not automatically open after clicking the above link, you can instead run the simulation by right clicking on the link and selecting ?save ta rget as?. after saving the file to your computer, start ltspice and open the demonstartion circuit by selecting "open" from the "file" menu. part number description price dc502a lt1767ems8e | 1.25mhz wide input range step down dc/dc converter, 3v to 25v input, 5v/3/3vout @ 1a $75.00 buy now back to top back to top back to top pa g e 5 of 5 linear technolo gy - lt1767 - monolithic 1.5a, 1.25mhz ste p -down switchin g re g ulat... 18-se p -2008 htt p ://www.linear.com/ p c/ p roductdetail. j s p ?navid=h0,c1,c1003,c1042,c1032,c1064,...

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